User’s Guide
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User’s Guide
1
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DYNAMIC WARNINGS AND RESTRICTIONS
It is important to operate this EVM within the input voltage range of 0 Vdc to100 Vdc.
Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM.
If there are questions concerning the input range, please contact a TI field representative prior to connecting
the input power.
Applying loads outside of the specified output range may result in unintended operation and/or possible
permanent damage to the EVM. Please consult the EVM User’s Guide prior to connecting any load to the EVM
output. If there is uncertainty as to the load specification, please contact a TI field representative.
During normal operation, some circuit components may have case temperatures greater than 50°C. The EVM
is designed to operate properly with certain components above 50°C as long as the input and output ranges are
maintained. These components include but are not limited to linear regulators, switching transistors, pass
transistors, and current sense resistors. These types of devices can be identified using the EVM schematic
located in the EVM User’s Guide. When placing measurement probes near these devices during operation,
please be aware that these devices may be very warm to the touch.
Mailing Address:
Texas Instruments
Post Office Box 655303
Dallas, Texas 75265
Copyright 2004, Texas Instruments Incorporated
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TPS40090 Multi-Phase Buck Converter and TPS2834
Drivers Steps-Down from 12-V to 1.5-V at 100 A
Systems Power
Contents
1
2
3
4
5
6
7
8
9
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Component Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Test Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Test Results/Performance Data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Layout Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
EVM Assembly Drawing and PCB Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
List of Materials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
1
Introduction
The TPS40090EVM−002 multi-phase dc-to-dc converter utilizes the TPS40090 multi-phase
controller and TPS2834 adaptive driver to step down a 12-V input to 1.5-V at 420 kHz. The
output current can exceed 100 A. The TPS40090 provides fixed-frequency, peak current-mode
control with forced-phase current balancing. Phase currents are sensed by the voltage drop
across the DC resistance (DCR) of inductors. Other features include a single voltage operation,
true differential output voltage sense, user programmable current limit, capacitor-programmable
soft-start and a power good indicator. Device operation is specified in the TPS40090
[1]
datasheet .
TPS40090EVM-002 can be configured into 2-, 3− or 4-phase operation. For 2-phase operation,
populate R65 and R66 to tie PWM2 and PWM4 up to internal 5-V and leave components in
related phases unpopulated. For 3-phase operation, tie PWM4 to BP5 through R66 only. For
4-phase operation, leave both R65 and R66 unpopulated.
In this user’s guide, all the tests are conducted under 4 phase operation.
4
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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2
Features
Table 1. TPS40090EVM−002 Performance Summary
PARAMETER
TEST CONDITIONS
MIN
10.5
TYP
12.0
MAX
14.0
UNITS
Input voltage range
V
A
Output voltage set point
Output current range
1.477
0
1.508
100
1.540
120
V
IN
= 12 V
I
rising from 10 A to 100 A,
OUT
(1)
Line regulation
0.1%
10.5 V ≤ V ≤ 14 V
IN
Load regulation
I
I
I
I
I
I
I
rising from 10 A to 100 A
rising from 10 A to 100 A
falling from 100 A to 10 A
rising from 10 A to 100 A
falling from 100 A to 10 A
0.3%
−160
200
< 10
< 15
89
OUT
OUT
OUT
OUT
OUT
OUT
OUT
Load transient response voltage
change
mV
PK
Load transient response recovery
time
µs
Loop bandwidth
= 100 A,
= 100 A
I
= 10 A
kHz
OUT
Phase margin
40
°
Input ripple voltage
Output ripple voltage
Output rise time
80
200
25
mV
PK
15
ms
Operating frequency
370
418
454
kHz
V
= 12 V,
= 100 A
V
V
= 1.5 V,
= 1.5 V,
IN
OUT
OUT
Full load efficiency
84.3%
I
OUT
V
= 12 V,
= 100 A
IN
Current sharing tolerance
5%
10%
I
OUT
3
Schematic
+
+
12V
Phase Programming
R65
R66
4−phase
3−phase
2−phase
open
open
1k
open
1k
1k
Figure 1. TPS40090EVM−002 Schematic Part 1 − TPS40090 Controller and Pre-Bias Circuit
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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TRANS_EN
Figure 2. TPS40090EVM−002 Schematic Part 2 − Driver Circuit and Load Transient Generator
6
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+
+
+
+
+
+
+
+
1.5V/100A
Figure 3. TPS40090EVM−002 Schematic Part 3 − Power Stage
4
Component Selection
4.1 Frequency of Operation
The clock oscillator frequency for the TPS40090 is programmed with a single resistor from RT
(pin 16) to signal ground. Equation (1) from the datasheet allows selection of the R resistor in
T
kΩ for a given switching frequency in kHz.
3
*1.024
PH
(1)
ǒ39.2 10 f
PH
* 7Ǔ (kW
)
R + R12 + K
T
where
•
K
is the coefficient that depends on the number of active phases
PH
•
•
•
f
is the single phase frequency, in kHz
PH
for 2-phase and 3-phase configurations K =1.333
PH
for 4-phase K =1.0 is a single phase frequency, kHz.
PH
The R resistor value is returned by the last expression in kΩ. For 420 kHz, R is calculated as
T
T
65.8 kΩ and a resistor with a 64.9-kΩ standard value is used.
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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4.2 Inductance Value
The output inductor value for each phase can be calculated from the volt-second during off time,
shown in equation (2).
V
V
OUT
OUT
(2)
L +
ǒ1 *
Ǔ
f I
V
RIPPLE
IN(max)
where
•
I
is usually chosen to be between 10% and 40% of maximum phase current I
.
PH(max)
RIPPLE
With I
= 20% of I
, there is a ripple current of 5 A, and the inductance value is found
RIPPLE
PH(max)
to be 0.63 µH. Using SPM12550−R62M300 inductors from TDK, each had inductance of 0.6µH
and resistance of 1.75-mΩ.
In multi-phase high current buck converter design, due to the ripple cancellation factor from
interleaving, the inductor value could be smaller than that in a single phase operation. But from
conduction loss point of view, the inductor value tends to be big to reduce the ripple current, thus
losses.
4.3 Input Capacitor Selection
The bulk input capacitor selection is based on the input voltage ripple requirements. Due to the
interleaving of multi phase, the input RMS current is reduced. The input ripple current RMS
value over load current is calculated in equation (3).
(3)
ǒ
Ǔ
D IIN(nom) NPH, D +
2
k ǒN , DǓ
k ǒN , DǓ ) 1
PH
ȱ
ȳ
ȴ
(
)
V
1 * D
N
PH
OUT
PH
) ǒ Ǔ
D *
* D
ȧǒ Ǔ ǒ Ǔȧ ƪ ƫ
2
N
N
PH
12 D
L f ǒIOUTǓ
PH
Ȳ
ȱ
3
3
ȳ
k ǒN , DǓ
k ǒN , DǓ ) 1
Ǔ2
k ǒN , DǓ ) 1
) k ǒN , DǓ2
PH
PH
ǒ
D *
* D
ǒ Ǔ ǒ Ǔȧ
ȧ
Ȳ
PH
PH
N
N
PH
PH
ȴ
where
kǒN , DǓ + floor N
ǒ
DǓ
PH
•
PH
•
•
floor(x) is the function to return the greatest integer less than N
× D
PH
N
is the number of active phases
PH
Figure 4 shows the input ripple current RMS value over the load current versus duty cycle with
different number of active phases.
8
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0.6
0.5
0.4
N
= 1
PH
0.3
0.2
0.1
N
= 2
PH
N
= 3
PH
N
= 4
80
PH
N
= 6
PH
0
0
10
20
30
40
50
60
70
90
100
Duty Cycle − %
Figure 4. Input Ripple Current RMS Value Overload Current
The maximum input ripple RMS current can be estimated as shown in (4).
(4)
ǒ
Ǔ
I ^ IOUT D IIN(nom) 4, Dmin + 3.18 A
It is also important to consider a minimum capacitance value which limits the voltage ripple to a
specified value if all the current is supplied by the onboard capacitor. For a typical ripple voltage
of 150 mV the maximum ESR is calculated in (5) as:
150 mV
3.18 A
D V
D I
(5)
ESR +
+
+ 47 mW
Two 68-µF, 20-V Oscon capacitors (20SVP68M) from Sanyo are placed on the input side of the
board. The ESR is 40 mΩ for each capacitor.
4.4 Output Ripple Cancellation and Capacitor Selection
Due to the interleaving of channels, the total output ripple current is smaller than the ripple
current from a single phase. The ripple cancellation factor is expressed in equation (6).
NPH
P
i + 1
Ťi * N
Ť
ǒ
D Ǔ
PH
OUT ǒN , DǓ +
(6)
DI
PH
*1
NPH
Ť
Ť
ǒ i * N D ) 1Ǔ
ƪ P
ƫ
PH
i + 1
k ǒN
Ǔ
OUT ǒN
(D), DI
Ǔ
ǒ
, D Ǔ
, D + if NPH v 1, DI
PH
OUT
PH
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where
•
•
•
D is the duty cycle for a single phase
is the number of active phases
N
PH
K (N ) is the intermediate function for calculation
PH
In this case, N =4 and D =0.107 which yields k=0.573.
PH
min
The actual output ripple is calculated in equation (7)
V
OUT
1.5 V
(7)
KǒN , DǓ +
I
+
0.573 + 3.41 A
RIPPLE
PH
L f
0.6m H 420 kHz
1.0
N
= 4
PH
0.8
0.6
N
= 3
PH
N
= 1
PH
0.4
0.2
0
N
= 2
PH
N
= 6
10
PH
0
20
30
40
50
60
70
80
90
100
Duty Cycle − %
Figure 5. Output Ripple Current Cancellation
Selection of the output capacitor is based on many application variables, including function, cost,
size, and availability. There are three ways to calculate the output capacitance.
1. The minimum allowable output capacitance is determined by the amount of inductor ripple
current and the allowable output ripple, as given in equation (8).
I
3.41 A
8 420 kHz 10 mV
RIPPLE
(8)
C
+
+
+ 101 mF
OUT(min)
8 f V
RIPPLE
In this design, C
is 101-µF with V
=10 mV. However, this affects only the
RIPPLE
OUT(min)
capacitive component of the ripple voltage, and the final value of capacitance is generally
influenced by ESR and transient considerations.
2. ESR limitation. (To limit the ripple voltage to 10 mV, the capacitor ESR should be less than
the value calculated in equation (9)).
V
10 mV
3.41 A
RIPPLE
RIPPLE
(9)
R t+
+
+ 2.93 mW
C
I
10
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3. Transient consideration. An additional consideration in the selection of the output inductor
and capacitance value can be derived from examining the transient voltage overshoot
which can be initiated with a load step from full load to no load. By equating the inductive
energy with the capacitive energy the equation (10) can be derived.
ǒ Ǔ2 ǒ Ǔ2
ǒ I
Ǔ
LEQ
* I
OL
0.6mH
2
OH
(
)
100 A
(10)
2
4
L I
V
+ ǒ(
Ǔ2 ǒ OUT1Ǔ2
) Ǔ
C
+
+
+ 1846 mF
OUT
2
2
2
)
(
1.75 V * 1.5 V
ǒV
* V
OUT2
where
•
•
•
•
I
I
is full load
is no load
OH
OL
V
V
is the the allowed transient voltage rise
is the initial voltage
OUT2
OUT1
In this 100-A design the capacitance required for limiting the transient is significantly larger than
the capacitance required to keep the ripple acceptably low. Eight 220-µF POSCAP capacitors
are in parallel with four 22-µF ceramic capacitors. The ESR of each POSCAP is 15mΩ.
4.5 MOSFET Selection
There are different requirements for switching FET(s) and rectifier FET(s) in the high-ratio step
down application. The duty cycle is around 12%. So the rectifier FET(s) is on for most of the
cycle. The conduction loss is dominant. Low-R
FET(s) are preferred. Also due to the dV/dt
DS(on)
turn on of the rectifier FET(s) and cross conduction, choose a rectifier FET with Qgs > Qgd.
When the switch node is falling, the Qgd can pull the gate of the lower FET below GND, which
upsets the driver. Two Si7880DP from Siliconix are in parallel for the rectifier FET. The R
this FET is 3 mΩ and Qgs=18nC, and Qgd=10.5nC.
of
DS(on)
The switching FET switches at high voltage and high current, the switching loss is dominant.
One single Si7860DP is selected for its low total gate charge.
Both types of FET(s) are offered in the Powerpak SO−8 package.
The PCB is layed out for two FETs in parallel, for both switching FET(s) and rectifier FET(s), to
give the feasibility to modify the board for different applications.
4.6 Current Sensing
TPS40090 supports both resistor current sensing and DCR current sensing approach. DCRs of
the output inductors are used in this design as the current sensing components. The DCR
current sensing circuit is shown in Figure 5. The idea is to parallel a R-C network to the inductor.
If the two time constants are same (L/DCR=R × C), then V =V
. Extra circuit, shown in (b), is
used to compensate the positive temperature coefficient of copper specific resistance, which is
C
DCR
0.385%/°C. See detail explanation in the datasheet.
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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With the chosen inductor described in Inductance Value, (section 4.2, of this document) the
following values are used.
•
•
•
•
•
R=19.6 kΩ
C=10 nF
R
=100 kΩ
NTC
R1=124 kΩ
R2=22.6 kΩ
L
DCR
V
IN
V
DCR
L
C
DCR
R
V
IN
V
OUT
R1
C
V
R
R2
C
R
NTC
R
THE
UDG−03136
Figure 6. DCR Current Sensing Circuit with Copper Temperature Compensation
4.7 Overcurrent Limit Protection
The overcurrent function monitors the voltage level separately on each current sense input and
compares it to the voltage on ILIM pin set by the divider from the controller’s reference.
If the threshold of V
/2.7 is exceeded, the PWM cycle on the respected phase is terminated.
ILIM
Voltage level on the ILIM pin is determined by (11).
+ 2.7 I R ; I + I
OUT
ǒV
2 L
Ǔ
* V
V
OUT
IN
OUT
(11)
V
)
ILIM
PH(max)
CS
PH(max)
f
V
SW
IN
OUT
where
•
•
I
is the maximum allowable value of the phase current
is the value of the current sense resistor
PH(max)
R
CS
12
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4.8 Compensation Components
The TPS40090 uses peak current mode control. Type II network is used here, which is
implemented to provide one zero and two poles. The first pole is placed at the origin to improve
DC regulation.
The ESR zero of the power stage is:
1
(12)
(13)
f
+
+ 354 kHz
ESRZ
2p R C
C
OUT
The zero is placed near 3.96 kHz to produce a reasonable time constant.
1
f +
Z
2p R11 C11
The second pole is placed at ESR zero (354 kHz).
1
f
+
P1
ǒ
Ǔ
Ǔ
(14)
C11 C12
ǒ Ǔ
2p R11
ǒ
C11)C12
The resulting values selected for this design are:
•
•
•
R11 = 40.2 kΩ
C11 = 1000 pF
C12 = 10 pF
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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5
Test Setup
The HPA072 has the following input/output connections: 12-V input J1 (VIN) and J2 (GND),
1.5-V output J9 (VOUT) and J10 (GND). A diagram showing the connection points is shown in
Figure 5. A power supply capable of supplying 18 A should be connected to VIN and GND
through a pair of 10 AWG wires. The 1.5-V load should be connected respectively to J9 and J10
through pairs of 0 AWG wires. Wire lengths should be minimized to reduce losses in the wires.
A 5-inch fan with 200-cfm air flow is recommended to operate this board at full load.
Oscilloscope
CH1
J8
J1
J2
J9
12 V/ 20 A
Power
Supply
TPS40090EVM−002
Board
Electronic
Load
J10
Fluke 45
DC
V
OUT
UDG−04063
Figure 7. Connections for the Test
14
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6
Test Results and Performance Data
6.1 Efficiency and Power Loss
Figure 8 shows the efficiency as the load varies from 10 A to over 100 A. The efficiency at full
load is about 84.3%.
Figure 7 shows the total loss versus the load current, which is approximately 28.3W at 100 A.
OVERALL EFFICIENCY
vs
OUTPUT CURRENT
TOTAL POWER LOSS
vs
OUTPUT CURRENT
40
35
90
85
V
SW
= 12 V
IN
V
SW
= 12 V
= 420 kHz
IN
f
= 420 kHz
f
30
25
80
20
15
75
70
10
5
0
65
0
20
40
60
80
100
120
0
20
40
60
80
100
120
I
− Output Current − A
OUT
I
− Output Current − A
OUT
Figure 9.
Figure 8.
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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6.2 Closed-Loop Performance
The TPS40090 uses peak current-mode control. Figure 10 shows the bode plots at 100 A of
load current, where no droop function is implemented. The crossover frequency is at 89 kHz with
phase margin of 40°.
GAIN AND PHASE
vs
OSCILLATOR FREQUENCY
80
60
180
135
90
PHASE
40
20
0
45
0
−45
−90
GAIN
−20
V
V
= 12 V
= 1.5 V
IN
OUT
= 10 A
−135
−180
I
OUT
−40
100
1 k
10 k
100 k
1 M
f
− Oscillator Frequency − kHz
OSC
Figure 10. Bode Plot
6.3 Output Ripple and Noise
Figure 11 shows typical output noise where V =12 V, and I
IN
than 10 mV.
=100A. The output ripple is less
OUT
I
= 100 A
OUT
Output Voltage Ripple
(10 mV/div)
t − Time − 500 ns / div
Figure 11. Output Noise
16
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6.4 Transient Response
The on-board load transient circuit enables to check the step load transient response on the
same board. Simply by putting a jumper to connect Pin1 and 2 of J3, a 90-A step load is created
by three 50-mΩ resistors placed on the board. The slew rates of the transient are 200 A/µs for
the load step-down and 160 A/µs for the load step-up.
The transient response is shown in Figure 6 as the load is stepped from 10 to 100 A. The output
deviation is approximately 200 mV and the settling time is within 15 µs.
Load Step = 90 A
Something Voltage
(10 mV/div)
Something Voltage
(10 mV/div)
t − Time − 20 µs / div
Figure 12. Transient Response
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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6.5 Start up with Pre-Biased Output
In synchronous buck converter, the bottom FET discharges the pre-biased output during
start-up. To avoid this, a comparator U9 and surround components are used to pull the SYNC
pin of the drivers low, which keeps the bottom FET off during startup. So the output can rise
smoothly. After the SS pin comes up, SYNC is pulled up high and enable the bottom FET’s
driving signal. The converter goes back to normal synchronization mode. This function can be
enabled by shorting J11 on the board.
Figure 8 shows the start-up waveform with pre−biased output with J11 short and open
respectively. In Figure 12, there are two glitches of SYNC waveform. The first one is cause by
P5V from TPS40090. When TPS40090 is enabled, P5V comes up first. SYNC is connected to
P5V through a divider. The second one happens when the driver is ready and turns on the
bottom FET when PWM signal is low. So the pre-biased output is pulled low which causes the
SYNC signal high to turn off the bottom FET. Then output voltage goes back and rises up
smoothly.
V
OUT
(2 V/div)
V
OUT
(2 V/div)
V
SYNC
(5 V/div)
V
SYNC
(2 V/div)
V
SS
(5 V/div)
V
SS
(5 V/div)
V
EN
(2 V/div)
V
EN
(2 V/div)
t − Time − 2.5 ms / div
t − Time − 1 ms / div
Figure 13. J11 Short Circuit
Figure 14. J11 Open Circuit
18
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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7
Layout Considerations
The PCB layout plays a critical role in the performance in a high frequency switching power
supply design. Following the suggestions listed below will help to improve the performance and
expedite the design.
•
To take full advantage of the ripple cancellation factor from interleaving, place the input
capacitors before the junction where the input voltage is distributed to each phase. Place the
output capacitors after the junction where all the inductors are connected;
•
Place the external drivers right next to the FETs and use at least 25 mil trace for gate drive
signal to improve noise immunity
•
•
•
•
•
•
Place some ceramic capacitors in the input of each channel to filter the current spikes
Place the NTC resistor right next to its related inductor for better thermal coupling
2 oz. or thicker copper is recommended to reduce the trace impedance
Place enough vias along pads of the power components to increase thermal conduction
Keep the current sensing traces as short as possible to avoid excessive noise pick up
Place the output inductors as symmetric as possible in relation to the output connectors to
obtain similar voltage drop from the trace impedance
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
19
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8
EVM Assembly Drawing and PCB Layout
Figure 15. Top Side Component Assembly
Figure 16. Bottom Assembly
20
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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Figure 17. Top Side Copper
Figure 18. Internal 1 (Ground Plane)
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
21
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Figure 19. Internal 2 (Power Plane)
Figure 20. Internal 3
22
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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Figure 21. Internal 4
Figure 22. Bottom Layer Copper
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
23
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SLUU195 − June 2004
9
List of Materials
The following table lists the TPS40090EVM−002 components corresponding to the schematic
shown in Figure 1.
Table 2. List of Materials
Reference
Designator
QTY
Description
Size
Manufacturer
Part Number
C1, C4
2
Capacitor, OS−CON, 68 µF, 20 V, 40 mΩ, 20%
10.3mm (F8)
Sanyo
20SVP68M
C2, C5, C7,
C8,C9, C10,
C11
7
Capacitor, ceramic, 1000−pF, 25 V, X7R, 5%
603
muRata
GRM39SL102J25
C3,C17, C18,
C19, C21
5
Capacitor, dielectric, 1.0 µF, 16 V, X7R, 10%
805
muRata
GRM40B105K16
C6
0
1
603
805
Std
Std
C12
Capacitor, ceramic, 0.01 µF, 50 V, X7R, 5%
muRata
GRM40UJ103J50
C13, C14, C15,
C16, C20, C22
GRM42−
65X5R475K16
6
Capacitor, dielectric, 4.7 µF, 16 V, X5R, 10%
1206
muRata
C30, C31, C32,
C33, C34, C35,
C36, C37
8
Capacitor, dielectric, 10 µF, 25 V, X5R
1210
TDK
C3225X5R1E106M
C38, C39, C40,
C41
4
4
Capacitor, ceramic, 1000−pF, 50 V, X7R, 5%
805
805
muRata
TDK
GRM40TH102J50
C42, C43, C44,
C45
Capacitor, ceramic, 0.01 µF, 50 V, COG
C2012COG1H103JT
C23, C24, C25,
C26,C46,
C47,C50, C51
8
Capacitor, POSCAP, 220 µF, 2.5 V, 15 mΩ, 20%
7343 (D)
Sanyo
2R5TPE220M
C48, C49, C52,
C53
4
5
Capacitor, Ceramic, 10 µF, 6.3 V, X5R
1206
TDK
C3216X5R0J106M
BAT54C
D1, D2, D3, D4,
D6
Diode, dual schottky, 200 mA, 30 V
SOT-23
Vishay−Liteon
D7, D8, D9,
D10
4
1
5
4
4
Diode, zener, 6.2 V, 350 mW
SOT−23
Copper
Diodes, Inc.
524600
BZX84C6V2
ILSCO
J1, J2, J9, J10
Lug, Solderless, #2 − #8 AWG, 1/4
Connector, shielded, test jack, vertical
Inductor, SMT, 0.62 µH, 30 A, 1.75 mΩ
MOSFET, N-channel, 30 V, 18 A, 8.0 mΩ
J4, J5, J6, J7,
J8
Johnson
Components
0.0125 DIA
129−0701−202
SPM12550−R62M300
Si7860DP
L1, L2, L3, L4
0.524 x 0.492
TDK
PWRPAK
S0−8
Q2, Q3, Q4, Q5
Vishay−Siliconix
PWRPAK
S0−8
Q6, Q7, Q8, Q9
0
8
MOSFET, N-channel, 30 V, 18 A, 8.0 mΩ
Vishay−Siliconix
Vishay−Siliconix
Si7860DP
Si7880DP
Q1, Q10, Q11,
Q12, Q13, Q14,
Q15, Q16
PWRPAK
S0−8
MOSFET, N-channel, 30 V, 29 A, 3 mΩ
R1
1
1
0
3
1
1
1
Resistor, chip, 8.25 kΩ, 1/16−W, 1%
Resistor, chip, 6.19 kΩ,, 1/16−W, 1%
603
603
603
603
603
603
603
Std
Std
Std
Std
Std
Std
Std
Std
Std
Std
Std
Std
Std
Std
R2
R3
R4, R9, R11
Resistor, chip, 10 kΩ, 1/16−W, 1%
Resistor, chip, 8.66 kΩ, 1/16−W, 1%
Resistor, chip, 49.9 Ω, 1/16−W, 1%
Resistor, chip, 40.2 kΩ, 1/16−W, 1%
R5
R6
R7
R8, R16, R55,
R56, R59, R60,
R61
7
Resistor, chip, 10−Ohms, 1/16−W, 1%
603
Std
Std
R10
R12
1
1
Resistor, chip, 475 kΩ, 1/16−W, 5%
Resistor, chip, 64.9 kΩ, 1/16−W, 1%
603
603
Std
Std
Std
Std
24
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
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Reference
Designator
QTY
Description
Resistor, chip, 2.2 Ω, 1/10−W, 1%
Resistor, chip, 19.6 kΩ, 1/10−W, 1%
Size
805
805
Manufacturer
Part Number
R27, R28, R29,
R30
4
4
Std
Std
Std
Std
R31, R32,R35,
R36
R33, R34, R37,
R38, R41, R42,
R45, R46
8
Resistor, chip, 2.7 Ω, 1/16−W, 1%
603
Std
Std
R39, R40, R43,
R44
4
4
4
Resistor, chip, 22.6 kΩ, 1/10−W, 1%
Resistor, chip, 124 kΩ, 1/10−W, 1%
NTC Resistor, chip, 100 kΩ, 1/10-W, 1%
805
805
805
Std
Std
Std
Std
R47, R49, R51,
R52
R48, R50, R53,
R54
NTHS0603N01N1003
J
Vishay
R65, R66
TP1
2
1
1
1
4
1
2
Resistor, chip, 1.0 kΩ, 1/10-W, 1%
Test point, 0.062 Hole, Red
805
0.25
Std
Keystone
Keystone
TI
Std
5011
TP2
Test point, 0.062 Hole, Black
0.25
5010
U1
IC, high-frequency, multiphase controller
IC, MOSFET driver, fast synchronous buck with DTC
IC, Precision timer
108,800
PWP14
TSSOP 8
0.038
TPS40090PW
TPS2834PWP
NE555PW
240−333
U2, U3, U4, U5
U6
TI
TI
E1, E2
Test point, black, 1 mm
Farnell
LOAD TRANSIENT CIRCUIT
PWRPAK
S0−8
Q17
1
0
3
0
MOSFET, N-channel, 12 V, 29 A, 3.0 mΩ,
MOSFET, N-channel, 12 V, 29 A, 3.0 mΩ,
Resistor, chip, 0.050 Ω,, 1-W, 0.5%
Resistor, chip, 0.050 Ω,, 1-W, 0.5%
Si7858DP
Si7858DP
PWRPAK
S0−8
Q18
WSL−2512−R050
0.5% R86
R18, R19, R21
R20, R22
2512
2512
Vishay
Vishay
WSL−2512−R050
0.5% R86
R23, R24
R25
2
1
1
2
1
1
1
1
Resistor, chip, 10 Ω, 1/16−W, 1%
Resistor, chip, 143 kΩ, 1/10−W, 1%
Resistor, chip, 1.43 kΩ, 1/10−W, 1%
Capacitor, ceramic, 0.1 µF, 25 V, X7R, 10%
Capacitor, ceramic, 0.01 µF, 50 V, X7R, 5%
Diode, dual ultra fast, series, 200 mA, 70 V
Diode, dual schottky, 200 mA, 30 V
603
805
Std
Std
Std
Std
R26
805
Std
Std
C27, C28
C29
805
TDK
C2012X7R1E104K
GRM40UJ103J50
BAV99
805
muRata
Fairchild
Vishay−Liteon
Sullins
D5
SOT23
SOT23
0.100 x 3
D6
BAT54C
J3
Header, 3-pin, 100 mil spacing, (36-pin strip)
PTC36SAAN
PRE-BIAS CIRCUIT
U7
1
IC, Single GP comparator, low voltage
SOT23−5
0.100 x 2
National
Sullins
LMV331M5
J11
1
Header, 2-pin, 100 mil spacing, (36-pin strip)
PTC36SAAN
R13, R15, R17,
R58
4
Resistor, chip, 10 kΩ, 1/16−W, 1%
603
Std
Std
R57
R14
1
1
Resistor, chip, 1 MΩ, 1/10−W, 1%
Resistor, chip, 8.66 kΩ, 1/16−W, 1%
805
603
Std
Std
Std
Std
TPS40090 Multi-Phase Buck Converter and TPS2834 Drivers Steps-Down from 12-V to 1.5-V at 100 A
25
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