Dual, Ultralow Noise
Variable Gain Amplifier
AD604
FUNCTIONAL BLOCK DIAGRAM
FEATURES
Ultralow input noise at maximum gain
0.80 nV/√Hz, 3.0 pA/√Hz
PAOx
–DSXx
+DSXx
VGNx
2 independent linear-in-dB channels
Absolute gain range per channel programmable
0 dB to 48 dB (preamplifier gain = 14 dB) through 6 dB to
54 dB (preamplifier gain = 20 dB)
1.0 dB gain accuracy
GAIN CONTROL
AND SCALING
VREF
DIFFERENTIAL
ATTENUATOR
R-1.5R
LADDER NETWORK
PAIx
OUTx
AFA
0dB TO –48.4dB
Bandwidth: 40 MHz (−3 dB)
VOCM
PROGRAMMABLE
ULTRALOW NOISE
PREAMPLIFIER
Input resistance: 300 kΩ
FIXED GAIN
AMPLIFIER
34.4dB
Variable gain scaling: 20 dB/V through 40 dB/V
Stable gain with temperature and supply variations
Single-ended unipolar gain control
Power shutdown at lower end of gain control
Drive ADCs directly
G = 14dB TO 20dB
PRECISION PASSIVE
INPUT ATTENUATOR
Figure 1.
APPLICATIONS
Ultrasound and sonar time-gain controls
High performance AGC systems
Signal measurement
GENERAL DESCRIPTION
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Preamplifier gains between 5 and 10 (14 dB and 20 dB) provide
overall gain ranges per channel of 0 dB through 48 dB and 6 dB
through 54 dB. The two channels of the AD604 can be cascaded
to provide greater levels of gain range by bypassing the preamplifier
of the second channel. However, in multiple channel systems,
cascading the AD604 with other devices in the AD60x VGA
family that do not include a preamplifier may provide a more
efficient solution. The AD604 provides access to the output of
the preamplifier, allowing for external filtering between the
preamplifier and the differential attenuator stage.
The AD604 is an ultralow noise, very accurate, dual-channel,
linear-in-dB variable gain amplifier (VGA) optimized for time-
based variable gain control in ultrasound applications; however,
it supports any application requiring low noise, wide bandwidth,
variable gain control. Each channel of the AD604 provides a
300 kΩ input resistance and unipolar gain control for ease of
use. User-determined gain ranges, gain scaling (dB/V), and dc
level shifting of output further optimize performance.
Each channel of the AD604 uses a high performance
preamplifier that provides an input-referred noise voltage of
0.8 nV/√Hz. The very accurate linear-in-dB response of the
AD604 is achieved with the differential input exponential
amplifier (DSX-AMP) architecture. Each DSX-AMP comprises
a variable attenuator of 0 dB to 48.36 dB followed by a high
speed fixed-gain amplifier. The attenuator is a 7-stage
R-1.5R ladder network. The attenuation between tap points is
6.908 dB and 48.36 dB for the ladder network.
Note that scale factors up to 40 dB/V are achievable with reduced
accuracy for scales above 30 dB/V. The gain scales linearly in
decibels with control voltages of 0.4 V to 2.4 V with the 20 dB/V
scale. Below and above this gain control range, the gain begins
to deviate from the ideal linear-in-dB control law. The gain
control region below 0.1 V is not used for gain control. When
the gain control voltage is <50 mV, the amplifier channel is
powered down to 1.9 mA.
The equation for the linear-in-dB gain response is
The AD604 is available in 24-lead SSOP, SOIC, and PDIP
packages and is guaranteed for operation over the −40°C to
+85°C temperature range.
G (dB) =
(Gain Scaling (dB/V) × VGN (V)) + (Preamp Gain (dB) – 19 dB)
Rev. E
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113 ©1996–2008 Analog Devices, Inc. All rights reserved.
AD604
SPECIFICATIONS
Each amplifier channel at TA = 25°C, VS = 5 V, RS = 50 Ω, RL = 500 Ω, CL = 5 pF, VREF = 2.50 V (scaling = 20 dB/V), 0 dB to 48 dB gain
range (preamplifier gain = 14 dB), VOCM = 2.5 V, C1 and C2 = 0.1 μF (see Figure 37), unless otherwise noted.
Table 1.
Parameter
Conditions
Min
Typ
Max
Unit
INPUT CHARACTERISTICS
Preamplifier
Input Resistance
Input Capacitance
Input Bias Current
Peak Input Voltage
300
8.5
−27
400
200
kΩ
pF
mA
mV
mV
Preamplifier gain = 14 dB
Preamplifier gain = 20 dB
VGN = 2.9 V, RS = 0 Ω
Input Voltage Noise
Preamplifier gain = 14 dB
Preamplifier gain = 20 dB
Independent of gain
RS = 50 Ω, f = 10 MHz, VGN = 2.9 V
RS = 200 Ω, f = 10 MHz, VGN = 2.9 V
0.8
0.73
3.0
2.3
1.1
nV/√Hz
nV/√Hz
pA/√Hz
dB
Input Current Noise
Noise Figure
dB
DSX
Input Resistance
Input Capacitance
Peak Input Voltage
Input Voltage Noise
Input Current Noise
Noise Figure
175
3.0
2.5
1.8
2.7
8.4
12
Ω
pF
V
nV/√Hz
pA/√Hz
dB
dB
dB
2
VGN = 2.9 V
VGN = 2.9 V
RS = 50 Ω, f = 10 MHz, VGN = 2.9 V
RS = 200 Ω, f = 10 MHz, VGN = 2.9 V
f = 1 MHz, VGN = 2.65 V
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Common-Mode Rejection Ratio
−20
OUTPUT CHARACTERISTICS
−3 dB Bandwidth
Slew Rate
Output Signal Range
Constant with gain
VGN = 1.5 V, output = 1 V step
RL ≥ 500 Ω
40
170
2.5 1.5
MHz
V/μs
V
Output Impedance
f = 10 MHz
2
Ω
Output Short-Circuit Current
40
mA
Harmonic Distortion
HD2
VGN = 1 V, VOUT = 1 V p-p
f = 1 MHz
−54
−67
−43
−48
dBc
dBc
dBc
dBc
HD3
HD2
HD3
f = 1 MHz
f = 10 MHz
f = 10 MHz
Two-Tone Intermodulation Distortion (IMD)
VGN = 2.9 V, VOUT = 1 V p-p
f = 1 MHz
−74
−71
−12.5
dBc
dBc
dBm
f = 10 MHz
Third-Order Intercept
f = 10 MHz, VGN = 2.65 V, VOUT = 1 V p-p,
input referred
1 dB Compression Point
Channel-to-Channel Crosstalk
f = 1 MHz, VGN = 2.9 V, output referred
VOUT = 1 V p-p, f = 1 MHz,
15
−30
dBm
dB
Channel 1: VGN = 2.65 V, inputs shorted,
Channel 2: VGN = 1.5 V (mid gain)
Group Delay Variation
VOCM Input Resistance
1 MHz < f < 10 MHz, full gain range
2
45
ns
kΩ
Rev. E | Page 3 of 32
AD604
Parameter
Conditions
Min
Typ
Max
Unit
ACCURACY
Absolute Gain Error
0 dB to 3 dB
3 dB to 43 dB
0.25 V < VGN < 0.400 V
0.400 V < VGN < 2.400 V
2.400 V < VGN < 2.65 V
0.400 V < VGN < 2.400 V
VREF = 2.500 V, VOCM = 2.500 V
VREF = 2.500 V, VOCM = 2.500 V
−1.2
−1.0
−3.5
+0.75
0.3
−1.25
0.25
30
+3
+1.0
+1.2
dB
dB
dB
dB/V
mV
mV
43 dB to 48 dB
Gain Scaling Error
Output Offset Voltage
Output Offset Variation
GAIN CONTROL INTERFACE
Gain Scaling Factor
−50
19
+50
50
30
VREF = 2.5 V, 0.4 V < VGN < 2.4 V
VREF = 1.67 V
Preamplifier gain = 14 dB
Preamplifier gain = 20 dB
20 dB/V, VREF = 2.5 V
20
30
21
dB/V
dB/V
dB
dB
V
μA
MΩ
μs
Gain Range
0 to 48
6 to 54
0.1 to 2.9
−0.4
2
Input Voltage (VGN) Range
Input Bias Current
Input Resistance
Response Time
48 dB gain change
0.2
VREF Input Resistance
POWER SUPPLY
10
kΩ
Specified Operating Range
One complete channel
One DSX only
5
5
V
V
Power Dissipation
One complete channel
One DSX only
VPOS, one complete channel
VPOS, one DSX only
VNEG, one preamplifier only
VPOS, VGN < 50 mV, one channel
VNEG, VGN < 50 mV, one channel
48 dB gain change, VOUT = 2 V p-p
220
95
32
19
−12
1.9
−150
0.6
mW
mW
mA
mA
mA
mA
μA
μs
Quiescent Supply Current
36
23
−15
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Powered Down
3.0
Power-Up Response Time
Power-Down Response Time
0.4
μs
Rev. E | Page 4 of 32
AD604
ABSOLUTE MAXIMUM RATINGS
Stresses above those listed under Absolute Maximum Ratings
Table 2.
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rating
Supply Voltage VS
Pin 17 to Pin 20 (with Pin 16, Pin 22 = 0 V)
Input Voltages
6.5 V
Pin 1, Pin 2, Pin 11, Pin 12
VPOS/2 2 V
continuous
Pin 4, Pin 9
Pin 5, Pin 8
Pin 6, Pin 7, Pin 13, Pin 14, Pin 23, Pin 24
Internal Power Dissipation
PDIP (N)
2 V
ESD CAUTION
VPOS, VNEG
VPOS, 0 V
2.2 W
SOIC (RW)
1.7 W
SSOP (RS)
1.1 W
Operating Temperature Range
Storage Temperature Range
Lead Temperature, Soldering 60 sec
−40°C to +85°C
−65°C to +150°C
300°C
θJA
AD604AN
AD604AR
AD604ARS
105°C/W
73°C/W
112°C/W
θJC
AD604AN
AD604AR
AD604ARS
35°C/W
38°C/W
34°C/W
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1 Pin 1, Pin 2, Pin 11 to Pin 14, Pin 23, and Pin 24 are part of a single-supply
circuit. The part is likely to suffer damage if any of these pins are accidentally
connected to VN.
2 When driven from an external low impedance source.
3 Using MIL-STD-883 test method G43-87 with a 1S (2-layer) test board.
Rev. E | Page 5 of 32
AD604
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
–DSX1
+DSX1
PAO1
FBK1
PAI1
1
2
3
4
5
6
7
8
9
24 VGN1
23 VREF
22 OUT1
21 GND1
20 VPOS
19 VNEG
18 VNEG
17 VPOS
16 GND2
15 OUT2
14 VOCM
13 VGN2
AD604
COM1
COM2
PAI2
TOP VIEW
(Not to Scale)
FBK2
PAO2 10
+DSX2 11
–DSX2 12
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
Mnemonic
–DSX1
+DSX1
PAO1
Description
1
2
3
Channel 1 Negative Signal Input to DSX1.
Channel 1 Positive Signal Input to DSX1.
Channel 1 Preamplifier Output.
4
FBK1
Channel 1 Preamplifier Feedback Pin.
5
PAI1
Channel 1 Preamplifier Positive Input.
6
7
8
COM1
COM2
PAI2
Channel 1 Signal Ground. When this pin is connected to positive supply, Preamplifier 1 shuts down.
Channel 2 Signal Ground. When this pin is connected to positive supply, Preamplifier 2 shuts down.
Channel 2 Preamplifier Positive Input.
9
FBK2
PAO2
+DSX2
–DSX2
VGN2
Channel 2 Preamplifier Feedback Pin.
Channel 2 Preamplifier Output.
Channel 2 Positive Signal Input to DSX2.
Channel 2 Negative Signal Input to DSX2.
Channel 2 Gain Control Input and Power-Down Pin. If this pin is grounded, the device is off; otherwise, positive
voltage increases gain.
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10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
VOCM
OUT2
GND2
VPOS
VNEG
VNEG
VPOS
GND1
OUT1
VREF
Input to this pin defines the common mode of the output at OUT1 and OUT2.
Channel 2 Signal Output.
Ground.
Positive Supply.
Negative Supply.
Negative Supply.
Positive Supply.
Ground.
Channel 1 Signal Output.
Input to this pin sets gain scaling for both channels to 2.5 V = 20 dB/V and 1.67 V = 30 dB/V.
Channel 1 Gain Control Input and Power-Down Pin. If this pin is grounded, the device is off; otherwise, positive
voltage increases gain.
VGN1
Rev. E | Page 6 of 32
AD604
TYPICAL PERFORMANCE CHARACTERISTICS
Unless otherwise noted, G (preamplifier) = 14 dB, VREF = 2.5 V (20 dB/V scaling), f = 1 MHz, RL = 500 Ω, CL = 5 pF, TA = 25°C, and
VSS
=
5 V.
50
40.0
37.5
35.0
32.5
30.0
27.5
25.0
22.5
20.0
40
30
20
10
0
THEORETICAL
3 CURVES
–40°C,
+25°C,
+85°C
ACTUAL
–10
0.1
0.5
0.9
1.3
1.7
2.1
2.5
2.9
1.25
1.50
1.75
2.00
2.25
2.50
VGN (V)
VREF (V)
Figure 3. Gain vs. VGN for Three Temperatures
Figure 6. Gain Scaling vs. VREF
2.0
1.5
1.0
0.5
60
50
G (PREAMP) = +14dB
(0dB TO +48dB)
40
G (PREAMP) = +20dB
(+6dB TO +54dB)
30
+25°C
–40°C
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0
20
–0.5
10
+85°C
DSX ONLY
(–14dB TO +34dB)
–1.0
0
–1.5
–10
–20
–2.0
0.2
0.7
1.2
1.7
2.2
2.7
0.1
0.5
0.9
1.3
1.7
2.1
2.5
2.9
VGN (V)
VGN (V)
Figure 7. Gain Error vs. VGN
Figure 4. Gain vs. VGN for Different Preamplifier Gains
50
2.0
1.5
1.0
ACTUAL
40
30
20
10
0
ACTUAL
30dB/V
VREF = 1.67V
0.5
FREQ = 1MHz
0
20dB/V
VREF = 2.5V
–0.5
–1.0
–1.5
–2.0
FREQ = 5MHz
FREQ = 10MHz
–10
0.1
0.2
0.7
1.2
1.7
2.2
2.7
0.5
0.9
1.3
1.7
2.1
2.5
2.9
VGN (V)
VGN (V)
Figure 8. Gain Error vs. VGN at Different Frequencies
Figure 5. Gain vs. VGN for Different Gain Scalings
Rev. E | Page 7 of 32
AD604
50
40
2.0
VGN = 2.5V
VGN = 2.9V
1.5
30
1.0
VGN = 1.5V
VGN = 0.5V
VGN = 0.1V
20
20dB/V
VREF = 2.5V
0.5
10
0
0
–10
–20
–30
–40
–50
–0.5
–1.0
–1.5
–2.0
30dB/V
VREF = 1.67V
VGN = 0V
0.2
0.7
1.2
1.7
2.2
2.7
100k
1M
10M
FREQUENCY (Hz)
100M
VGN (V)
Figure 9. Gain Error vs. VGN for Two Gain Scaling Values
Figure 12. AC Response for Various Values of VGN
25
2.55
2.54
2.53
2.52
2.51
2.50
2.49
2.48
2.47
N = 50
VOCM = 2.5V
VGN1 = 1.0V
VGN2 = 1.0V
ΔG(dB) =
20
–40°C
G(CH1) – G(CH2)
15
10
5
+25°C
+85°C
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2.46
0
2.45
0.2
–1.0 –0.8 –0.6 –0.4 –0.2
0.1
0.3
0.5
0.7
0.9
0.7
1.2
1.7
2.2
2.7
DELTA GAIN (dB)
VGN (V)
Figure 10. Gain Match; VGN1 = VGN2 = 1.0 V
Figure 13. Output Offset vs. VGN for Three Temperatures
25
20
15
10
5
210
N = 50
VGN1 = 2.50V
VGN2 = 2.50V
ΔG(dB) =
190
170
150
130
110
90
G(CH1) – G(CH2)
+85°C
+25°C
1.7
–40°C
0
–1.0 –0.8 –0.6 –0.4 –0.2
0.1
0.3
0.5
0.7
0.9
0.1
0.5
0.9
1.3
VGN (V)
2.1
2.5
2.9
DELTA GAIN (dB)
Figure 11. Gain Match; VGN1 = VGN2 = 2.50 V
Figure 14. Output Referred Noise vs. VGN for Three Temperatures
Rev. E | Page 8 of 32
AD604
1000
100
10
10
VGN = 2.9V
1
R
ALONE
1
SOURCE
0.1
0.1
0.1
0.5
0.9
1.3
1.7
2.1
2.5
2.9
1
10
100
1k
VGN (V)
R
(Ω)
SOURCE
Figure 18. Input Referred Noise vs. RSOURCE
Figure 15. Input Referred Noise vs. VGN
16
15
14
13
12
11
10
9
900
850
800
750
700
650
600
VGN = 2.9V
VGN = 2.9V
8
7
6
5
4
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3
2
1
1
10
100
1k
10k
–40
–20
0
20
40
60
80 90
R
(Ω)
TEMPERATURE (°C)
SOURCE
Figure 16. Input Referred Noise vs. Temperature
Figure 19. Noise Figure vs. RSOURCE
40
35
30
25
20
15
10
5
770
765
760
755
750
745
740
VGN = 2.9V
R
= 240Ω
S
0
100k
1M
10M
0
0.4
0.8
1.2
1.6
2.0
2.4
2.8
FREQUENCY (Hz)
VGN (V)
Figure 17. Input Referred Noise vs. Frequency
Figure 20. Noise Figure vs. VGN
Rev. E | Page 9 of 32
AD604
–20
–30
–40
–45
–50
–55
–60
–65
–70
V
= 1V p-p
V
= 1V p-p
O
O
VGN = 1V
VGN = 1V
–40
–50
–60
HD2
–70
–80
HD3
–90
–100
–110
–120
9.96
9.98
10.00
10.02
10.04
100k
1M
10M
FREQUENCY (Hz)
100M
FREQUENCY (MHz)
Figure 24. Intermodulation Distortion
Figure 21. Harmonic Distortion vs. Frequency
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
5
0
V
= 1V p-p
O
INPUT
HD2 (10MHz)
SIGNAL
LIMIT
–5
800mV p-p
–10
–15
–20
–25
10MHz
HD3 (10MHz)
1MHz
HD2 (1MHz)
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–30
HD3 (1MHz)
–35
0.1
0.5
0.9
1.3
1.7
2.1
2.5
2.9
0.5
0.9
1.3
1.7
2.1
2.5
2.9
VGN (V)
VGN (V)
Figure 22. Harmonic Distortion vs. VGN
Figure 25. 1 dB Compression vs. VGN
–20
–30
–40
–50
–60
–70
–80
25
V
= 1V p-p
R
S
V
= 1V p-p
O
O
VGN = 1V
DUT
20
15
10
5
50Ω
500Ω
f
= 1MHz
HD2 (10MHz)
HD3 (10MHz)
HD2 (1MHz)
0
f
= 10MHz
HD3 (1MHz)
–5
–10
–15
0
50
100
R
150
200
250
0.4
0.9
1.4
1.9
2.4
2.9
(Ω)
VGN (V)
SOURCE
Figure 26. Third-Order Intercept vs. VGN
Figure 23. Harmonic Distortion vs. RSOURCE
Rev. E | Page 10 of 32
AD604
2V
V
= 2V p-p
O
VGN = 1.5V
500mV
2.9V 100
90
10
0%
0.1V
500mV
100ns
–2V
253ns
100ns/DIV
1.253µs
Figure 27. Large Signal Pulse Response
Figure 30. Gain Response
0
200
VGN1 = 1V
V
= 200mV p-p
O
V
V
= 1V p-p
VGN = 1.5V
OUT1
= GND
–10
–20
–30
–40
–50
–60
IN2
VGN2 = 2.9V
VGN2 = 2V
TRIG'D
–200
VGN2 = 1.5V
VGN2 = 0.1V
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–70
100k
1M
10M
FREQUENCY (Hz)
100M
253ns
100ns/DIV
1.253µs
Figure 28. Small Signal Pulse Response
Figure 31. Crosstalk (Channel 1 to Channel 2) vs. Frequency
0
500mV
–10
2.9V
100
90
VGN = 2.9V
–20
VGN = 2.5V
–30
VGN = 2V
–40
10
0%
0V
–50
VGN = 0.1V
500mV
200ns
–60
100k
1M
10M
FREQUENCY (Hz)
100M
Figure 32. DSX Common-Mode Rejection Ratio vs. Frequency
Figure 29. Power-Up/Power-Down Response
Rev. E | Page 11 of 32
AD604
1M
100k
10k
1k
40
35
30
25
20
15
10
5
+I (AD604) = +I (PA) + +I (DSX)
S
S
S
–I (AD604) = –I (PA)
S
S
AD604 (+I
)
S
DSX (+I )
S
100
10
PREAMP (±I )
S
+I (VGN = 0)
S
0
–40
1
1k
–20
0
20
40
60
80 90
10k
100k
1M
10M
100M
TEMPERATURE (°C)
FREQUENCY (Hz)
Figure 33. Input Impedance vs. Frequency
Figure 35. Supply Current (One Channel) vs. Temperature
27.6
27.4
27.2
27.0
26.8
26.6
26.4
26.2
26.0
25.8
20
18
16
14
VGN = 0.1V
12
10
VGN = 2.9V
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8
6
100k
–40
–20
0
20
40
60
80 90
1M
10M
100M
TEMPERATURE (°C)
FREQUENCY (Hz)
Figure 34. Input Bias Current vs. Temperature
Figure 36. Group Delay vs. Frequency
Rev. E | Page 12 of 32
AD604
THEORY OF OPERATION
The AD604 is a dual-channel VGA with an ultralow noise
one channel. Each identical channel consists of a preamplifier
with gain setting resistors (R5, R6, and R7) and a single-supply
X-AMP® (hereafter called DSX, differential single-supply X-AMP)
made up of the following:
example, if the preamp gain is set to 14 dB and VREF is set to
2.50 V (to establish a gain scaling of 20 dB/V), the gain equation
simplifies to
G (dB) = 20 (dB/V) × VGN (V) – 5 dB
The desired gain can then be achieved by setting the unipolar
gain control (VGN) to a voltage within its nominal operating
range of 0.25 V to 2.65 V (for 20 dB/V gain scaling). The gain is
monotonic for a complete gain control voltage range of 0.1 V to
2.9 V. Maximum gain can be achieved at a VGN of 2.9 V.
•
•
•
A precision passive attenuator (differential ladder).
A gain control block.
A VOCM buffer with supply splitting resistors
(R3 and R4).
The inputs VREF and VOCM are common to both channels.
They are decoupled to ground, minimizing interchannel
crosstalk. For the highest gain scaling accuracy, VREF should
have an external low impedance voltage source. For low accuracy
20 dB/V applications, the VREF input can be decoupled with a
capacitor to ground. In this mode, the gain scaling is determined
by the midpoint between VPOS and GND; therefore, care
should be taken to control the supply voltage to 5 V. The input
resistance looking into the VREF pin is 10 kΩ 20%.
•
An active feedback amplifier (AFA) with gain setting
resistors (R1 and R2). To understand the active-feedback
AD830 is a practical implementation of the idea.
The preamplifier is powered by a 5 V supply, while the DSX
uses a single +5 V supply. The linear-in-dB gain response of the
AD604 can generally be described by
G (dB) = Gain Scaling (dB/V) × Gain Control (V) +
(Preamp Gain (dB) − 19 dB)
The DSX portion of the AD604 is a single-supply circuit, and
the VOCM pin is used to establish the dc level of the midpoint
of this portion of the circuit. The VOCM pin only needs an
external decoupling capacitor to ground to center the midpoint
between the supply voltages (5 V, GND); however, the VOCM
can be adjusted to other voltage levels if the dc common-mode
(1)
Each channel provides between 0 dB to 48.4 dB and 6 dB to 54.4
dB of gain, depending on the user-determined preamplifier
gain. The center 40 dB of gain is exactly linear-in-dB while the
gain error increases at the top and bottom of the range. The gain
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level of the output is important to the user (for example, see the
of the preamplifier is typically either 14 dB or 20 dB but can be
section entitled Medical Ultrasound TGC Driving the AD9050,
set to intermediate values by a single external resistor (see the
a 10-Bit, 40 MSPS ADC). The input resistance looking into the
VOCM pin is 45 kΩ 20%.
Preamplifier section for details). The gain of the DSX can vary
from −14 dB to +34.4 dB, as determined by the gain control
voltage (VGN). The VREF input establishes the gain scaling;
the useful gain scaling range is between 20 dB/V and 40 dB/V
for a VREF voltage of 2.5 V and 1.25 V, respectively. For
VREF
VGNx
GAIN
CONTROL
175Ω
C1
C2
PAIx
PAOx
EXT.
+DSXx
–DSXx
DISTRIBUTED G
Ao
M
R7
DIFFERENTIAL
ATTENUATOR
G1
G2
40Ω
FBKx
R5
32Ω
OUTx
175Ω
R6
VPOS
R3
8Ω
COMx
200kΩ
R2
20Ω
R1
820Ω
VOCM
C3
EXT.
R4
200kΩ
Figure 37. Simplified Block Diagram of a Single Channel of the AD604
Rev. E | Page 13 of 32
AD604
preamplifier to be 17.7 dB. The −3 dB small signal bandwidth of
one complete channel of the AD604 (preamplifier and DSX) is
40 MHz and is independent of gain.
PREAMPLIFIER
The input capability of the following single-supply DSX (2.5 2 V
for a +5 V supply) limits the maximum input voltage of the
preamplifier to 400 mV for the 14 dB gain configuration or
200 mV for the 20 dB gain configuration.
To achieve optimum specifications, power and ground manage-
ment are critical to the AD604. Large dynamic currents result
because of the low resistances needed for the desired noise
performance. Most of the difficulty is with the very low gain
setting resistors of the preamplifier that allow for a total input
referred noise, including the DSX, as low as 0.8 nV/√Hz. The
consequently large dynamic currents have to be carefully
handled to maintain performance even at large signal levels.
20
The preamplifier gain can be programmed to 14 dB or 20 dB by
either shorting the FBK1 node to PAO1 (14 dB) or by leaving
the FBK1 node open (20 dB). These two gain settings are very
accurate because they are set by the ratio of the on-chip resistors.
Any intermediate gain can be achieved by connecting the
appropriate resistor value between PAO1 and FBK1 according
to Equation 2 and Equation 3.
OPEN
19
(
R7 || REXT + R5 + R6
)
VOUT
VIN
40Ω
18
17
16
15
14
13
12
11
10
G =
=
(2)
(3)
R6
[
R6 × G −
(
R5 + R6 × R7
)
]
REXT
=
R7 −
(
R6 × G
)
+
(R5 + R6)
SHORT
Because the internal resistors have an absolute tolerance of 20%,
the gain can be in error by as much as 0.33 dB when REXT is 30 Ω,
where it is assumed that REXT is exact.
IN
V
IN
150Ω
50Ω
40Ω
R
EXT
8Ω
32Ω
Figure 38 shows how the preamplifier is set to gains of 14 dB,
17.5 dB, and 20 dB. The gain range of a single channel of the
AD604 is 0 dB to 48 dB when the preamplifier is set to 14 dB
100k
1M
10M
FREQUENCY (Hz)
100M
Figure 39. AC Response for Preamplifier Gains of 14 dB, 17.5 dB, and 20 dB
The preamplifier uses a dual 5 V supply to accommodate large
dynamic currents and a ground referenced input. The preamplifier
PAI1
output is also ground referenced and requires a common-mode
level shift into the single-supply DSX. The two external coupling
+DSXx, and –DSXx, nodes and ground, respectively, perform
eliminate any offset that would otherwise be introduced by the
preamplifier. It should be noted that an offset of 1 mV at the
input of the DSX is amplified by 34.4 dB (× 52.5) when the gain
control voltage is at its maximum; this equates to 52.5 mV at the
output. AC coupling is consequently required to keep the offset
from degrading the output signal range.
PAO1
R7
40Ω
R6
8Ω
R5
32Ω
COM1
FBK1
a. PREAMP GAIN = 14dB
PAI1
PAO1
R7
40Ω
R10
40Ω
R6
8Ω
R5
32Ω
COM1
FBK1
b. PREAMP GAIN = 17.5dB
The gain-setting preamplifier feedback resistors are small
enough (8 Ω and 32 Ω) that even an additional 1 Ω in the
ground connection at Pin COM1 (the input common-mode
reference) seriously degrades gain accuracy and noise performance.
This node is sensitive, and careful attention is necessary to
minimize the ground impedance. All connections to the COM1
node should be as short as possible.
PAI1
PAO1
R7
40Ω
R6
8Ω
R5
32Ω
COM1
FBK1
c. PREAMP GAIN = 20dB
Figure 38. Preamplifier Gain Programmability
The preamplifier, including the gain setting resistors, has a
noise performance of 0.71 nV/√Hz and 3 pA/√Hz. Note that a
significant portion of the total input referred voltage noise is
due to the feedback resistors. The equivalent noise resistance
presented by R5 and R6 in parallel is nominally 6.4 Ω, which
contributes 0.33 nV/√Hz to the total input referred voltage noise.
For a preamplifier gain of 14 dB, the −3 dB small signal bandwidth
of the preamplifier is 130 MHz. When the gain is at its maximum
of 20 dB, the bandwidth is reduced by half to 65 MHz. Figure 39
shows the ac response for the three preamp gains shown in
17.5 dB, but the mismatch between the internal resistors and
the external resistor causes the actual gain for this particular
Rev. E | Page 14 of 32
AD604
The larger portion of the input referred voltage noise comes
from the amplifier with 0.63 nV/√Hz. The current noise is
independent of gain and depends only on the bias current in
the input stage of the preamplifier, which is 3 pA/√Hz.
A unique circuit technique is used to interpolate continuously
among the tap points, thereby providing continuous attenuation
from 0 dB to −48.36 dB. The ladder network, together with the
interpolation mechanism, can be considered a voltage-controlled
potentiometer.
The preamplifier can drive 40 Ω (the nominal feedback resistors)
and the following 175 Ω ladder load of the DSX with low
distortion. For example, at 10 MHz and 1 V at the output, the
preamplifier has less than −45 dB of second and third harmonic
distortion when driven from a low (25 Ω) source resistance.
Because the DSX circuit uses a single voltage power supply, the
input biasing is provided by the VOCM buffer driving the MID
have to dc bias the inputs externally. If not done carefully, the
biasing network can introduce additional noise and offsets. By
providing internal biasing, the user is relieved of this task and
only needs to ac-couple the signal into the DSX. Note that the
input to the DSX is still fully differential if driven differentially;
that is, Pin +DSXx and Pin −DSXx see the same signal but with
opposite polarity (see the Ultralow Noise, Differential Input-
Differential Output VGA section).
In applications that require more than 48 dB of gain range, two
AD604 channels can be cascaded. Because the preamplifier has
a limited input signal range and consumes over half (120 mW)
of the total power (220 mW), and its ultralow noise is not necessary
after the first AD604 channel, a shutdown mechanism that
disables only the preamplifier is provided. To shut down the
preamplifier, connect the COM1 pin and/or COM2 pin to the
positive supply; the DSX is unaffected. For additional details,
What changes is the load seen by the driver; it is 175 Ω when
each input is driven single-ended but 350 Ω when driven
differentially. This is easily explained by thinking of the ladder
network as two 175 Ω resistors connected back-to-back with
the middle node, MID, being biased by the VOCM buffer. A
differential signal applied between the +DSXx and −DSXx
nodes results in zero current into the MID node, but a single-
ended signal applied to either input, +DSXx or –DSXx, while
the other input is ac-grounded causes the current delivered by
the source to flow into the VOCM buffer via the MID node.
1
2
3
4
5
6
7
8
9
–DSX1
+DSX1
PAO1
FBK1
PAI1
24
23
22
21
20
19
18
17
16
VGN1
VREF
OUT1
GND1
VPOS
COM1AD604VNEG
COM2
PAI2
VNEG
VPOS
GND2
FBK2
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The ladder resistor value of 175 Ω provides the optimum
10
11
12
15
14
13
PAO2
OUT2
VOCM
VGN2
balance between the load driving capability of the preamplifier
and the noise contribution of the resistors. An advantage of the
X-AMP architecture is that the output referred noise is constant
tap resistance is equal for all taps after only a few taps away
from the inputs. The resistance seen looking into each tap is
54.4 Ω, which makes 0.95 nV/√Hz of Johnson noise spectral
density. Because there are two attenuators, the overall noise
contribution of the ladder network is √2 times 0.95 nV/√Hz
or 1.34 nV/√Hz, a large fraction of the total DSX noise. The
balance of the DSX circuit components contributes another
1.2 nV/√Hz, which together with the attenuator produces
1.8 nV/√Hz of total DSX input referred noise.
+DSX2
–DSX2
Figure 40. Shutdown of Preamplifiers Only
DIFFERENTIAL LADDER (ATTENUATOR)
The attenuator before the fixed-gain amplifier of the DSX is
realized by a differential 7-stage R-1.5R resistive ladder network
with an untrimmed input resistance of 175 Ω single-ended or
350 Ω differential. The signal applied at the input of the ladder
network is attenuated by 6.908 dB per tap; thus, the attenuation
at the first tap is 0 dB, at the second, 13.816 dB, and so on, all
the way to the last tap where the attenuation is 48.356 dB
–6.908dB
R
R
–13.82dB
R
R
–20.72dB
R
R
–27.63dB
R
R
–34.54dB
R
R
–41.45dB
R
R
–48.36dB
R
+DSXx
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
175Ω
MID
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
175Ω
R
–DSXx
NOTES
1. R = 96Ω
2. 1.5R = 144Ω
Figure 41. R-1.5R Dual Ladder Network
Rev. E | Page 15 of 32
AD604
From these equations, it can be seen that all gain curves intercept at
the same −5 dB point; this intercept is +6 dB higher (+1 dB) if
the preamplifier gain is set to +20 dB or +14 dB lower (−19 dB)
if the preamplifier is not used at all. Outside the central linear
range, the gain starts to deviate from the ideal control law but
still provides another 8.4 dB of range. For a given gain scaling,
AC COUPLING
The DSX portion of the AD604 is a single-supply circuit and,
therefore, its inputs need to be ac-coupled to accommodate
ground-based signals. External Capacitors C1 and C2 in Figure 37
level shift the ground referenced preamplifier output from
ground to the dc value established by VOCM (nominal 2.5 V).
C1 and C2, together with the 175 Ω looking into each of the
DSX inputs (+DSXx and −DSXx), act as high-pass filters with
corner frequencies depending on the values chosen for C1 and
C2. As an example, for values of 0.1 μF at C1 and C2, combined
with the 175 Ω input resistance at each side of the differential
ladder of the DSX, the −3 dB high-pass corner is 9.1 kHz.
VREF can be calculated as shown in Equation 7.
2.500 V × 20 dB/V
VREF =
(7)
Gain Scale
Usable gain control voltage ranges are 0.1 V to 2.9 V for the
20 dB/V scale and 0.1 V to 1.45 V for the 40 dB/V scale. VGN
voltages of less than 0.1 V are not used for gain control because
below 50 mV the channel (preamplifier and DSX) is powered
down. This can be used to conserve power and, at the same
time, to gate off the signal. The supply current for a powered-
down channel is 1.9 mA; the response time to power the device
on or off is less than 1 μs.
If the AD604 output needs to be ground referenced, another
ac coupling capacitor is required for level shifting. This
capacitor also eliminates any dc offsets contributed by the DSX.
With a nominal load of 500 Ω and a 0.1 μF coupling capacitor,
this adds a high-pass filter with −3 dB corner frequency at about
3.2 kHz.
ACTIVE FEEDBACK AMPLIFIER (FIXED-GAIN AMP)
The choice for all three of these coupling capacitors depends on
the application. They should allow the signals of interest to pass
unattenuated while, at the same time, they can be used to limit
the low frequency noise in the system.
To achieve single-supply operation and a fully differential input
to the DSX, an active feedback amplifier (AFA) is used. The
AFA is an op amp with two gm stages; one of the active stages is
used in the feedback path (therefore the name), while the other
is used as a differential input. Note that the differential input is
an open-loop gm stage that requires it to be highly linear over
the expected input signal range. In this design, the gm stage that
senses the voltages on the attenuator is a distributed one; for
GAIN CONTROL INTERFACE
The gain control interface provides an input resistance of
approximately 2 MΩ at VGN1 and gain scaling factors from
20 dB/V to 40 dB/V for VREF input voltages of 2.5 V to 1.25 V,
respectively. The gain scales linearly in decibels for the center 40
dB of gain range, which for VGN is equal to 0.4 V to 2.4 V for
the 20 dB/V scale and 0.2 V to 1.2 V for the 40 dB/V scale. Figure
42 shows the ideal gain curves for a nominal preamplifier gain
of 14 dB, which are described by the following equations:
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example, there are as many gm stages as there are taps on the
ladder network. Only a few of them are on at any one time,
depending on the gain control voltage.
The AFA makes a differential input structure possible because
one of its inputs (G1) is fully differential; this input is made up
of a distributed gm stage. The second input (G2) is used for
feedback. The output of G1 is some function of the voltages
sensed on the attenuator taps, which is applied to a high-gain
amplifier (A0). Because of negative feedback, the differential
input to the high-gain amplifier has to be zero; this in turn
implies that the differential input voltage to G2 times gm2 (the
transconductance of G2) has to be equal to the differential
input voltage to G1 times gm1 (the transconductance of G1).
G (20 dB/V) = 20 × VGN – 5, VREF = 2.500 V
G (20 dB/V) = 30 × VGN – 5, VREF = 1.666 V
G (20 dB/V) = 40 × VGN – 5, VREF = 1.250 V
(4)
(5)
(6)
50
45
40
35
40dB/V
30dB/V
20dB/V
Therefore, the overall gain function of the AFA is
30
25
20
15
10
5
VOUT
g m1 R1 + R2
=
×
(8)
LINEAR-IN-dB RANGE
OF AD604 WITH
PREAMPLIFIER
SET TO 14dB
VATTEN
g m2
R2
where:
V
V
OUT is the output voltage.
ATTEN is the effective voltage sensed on the attenuator.
0
0.5
1.0
1.5
2.0
2.5
3.0
(R1 + R2)/R2 = 42
gm1/gm2 = 1.25
–5
GAIN CONTROL VOLTAGE (VGN)
Figure 42. Ideal Gain Curves vs. VGN
The overall gain is thus 52.5 (34.4 dB).
Rev. E | Page 16 of 32
AD604
The AFA offers the following additional features:
Under normal operating conditions, it is best to connect a
decoupling capacitor to VOCM, in which case, the common-
mode voltage of the DSX is half the supply voltage, which allows
for maximum signal swing. Nevertheless, the common-mode
voltage can be shifted up or down by directly applying a voltage
to VOCM. It can also be used as another signal input, the only
limitation being the rather low slew rate of the VOCM buffer.
•
•
•
The ability to invert the signal by switching the positive
and negative inputs to the ladder network
The possibility of using DSX1 input as a second signal
input
Fully differential high-impedance inputs when both
preamplifiers are used with one DSX (the other DSX could
still be used alone)
If the dc level of the output signal is not critical, another coupling
capacitor is normally used at the output of the DSX; again, this
is done for level shifting and to eliminate any dc offsets contributed
•
Independent control of the DSX common-mode voltage
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Rev. E | Page 17 of 32
AD604
APPLICATIONS INFORMATION
channel of the AD604. The signal is applied at Pin 5. RGN is
normally 0, in which case the preamplifier is set to a gain of 5
(14 dB). When FBK1 is left open, the preamplifier is set to a
gain of 10 (20 dB), and the gain range shifts up by 6 dB. The ac
coupling capacitors before −DSX1 and +DSX1 should be selected
according to the required lower cutoff frequency. In this example,
the 0.1 μF capacitors, together with the 175 Ω seen looking into
each of the DSXx input pins, provide a −3 dB high-pass corner
of about 9.1 kHz. The upper cutoff frequency is determined by
the bandwidth of the channel, which is 40 MHz. Note that the
signal can be simply inverted by connecting the output of the
preamplifier to −DSX1 instead of +DSX1; this is due to the fully
differential input of the DSX.
VREF requires a voltage of 1.25 V to 2.5 V, with between 40 dB/V
and 20 dB/V gain scaling, respectively. Voltage VGN controls
the gain; its nominal operating range is from 0.25 V to 2.65 V
for 20 dB/V gain scaling and 0.125 V to 1.325 V for 40 dB/V
scaling. When VGNx is grounded, the channel powers down
and disables its output.
COM1 is the main signal ground for the preamplifier and needs
to be connected with as short a connection as possible to the input
ground. Because the internal feedback resistors of the preamplifier
are very small for noise reasons (8 Ω and 32 Ω nominally), it is
of utmost importance to keep the resistance in this connection
to a minimum. Furthermore, excessive inductance in this
connection can lead to oscillations.
Because of the ultralow noise and wide bandwidth of the
AD604, large dynamic currents flow to and from the power
supply. To ensure the stability of the part, careful attention to
supply decoupling is required. A large storage capacitor in
parallel with a smaller high-frequency capacitor connected at
the supply pins, together with a ferrite bead coming from the
supply, should be used to ensure high-frequency stability.
0.1µF
1
24
23
22
21
20
19
18
VGN
–DSX1
+DSX1
PAO1
VGN1
VREF
OUT1
2
3
+2.5V
0.1µF
0.1µF
OUT
RGN
4
FBK1AD604 GND1
R
L
V
5
IN
PAI1
VPOS
VNEG
VNEG
500Ω
+5V
–5V
6
COM1
COM2
PAI2
7
8
To provide for additional flexibility, COM1 can be used to
disable the preamplifier. When COM1 is connected to VP, the
preamplifier is off, yet the DSX portion can be used independently.
This may be of value when cascading the two DSX stages in the
AD604. In this case, the first DSX output signal with respect to
VPOS 17
9
16
15
14
13
FBK2
PAO2
+DSX2
–DSX2
GND2
OUT2
VOCM
VGN2
10
11
12
0.1µF
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noise is large and using the second preamplifier at this point
Figure 43. Basic Connections for a Single Channel
eliminated if VOCM is biased at the same 3.3 V common-mode
Rev. E | Page 18 of 32
AD604
C1
0.1µF
1
2
–DSX1
+DSX1
VGN1 24
VREF 23
OUT1 22
VREF
C2
0.1µF
AD604
3
PAO1
FBK1
PAI1
VSET (<0V)
4
21
20
19
18
17
16
15
14
13
GND1
VPOS
VNEG
VNEG
VPOS
GND2
OUT2
VOCM
VGN2
5
R8
VIN
(MAX
800mV p-p)
+5V
–5V
–5V
+5V
C11
1µF
R1
49.9Ω
2kΩ
2
– (V1)
1V
6
COM1
COM2
PAI2
R4
C8
LOW-
PASS
2kΩ
0.33µF
7
OFFS
NULL
FILTER
1
2
3
8
7
6
5
NC
+V
+5V
8
+5V
V1 = V × G
IN
R7
1kΩ
S
8
7
6
5
AD711
9
FBK2
PAO2
+DSX2
–DSX2
C7
0.33µF
X1
X2 VP
AD835
W
OUT
VG
C10
1µF
10
11
12
C3
0.1µF
OFFS
NULL
4
–5V
–V
S
C7
0.1µF
R3
C6
1kΩ
Y1 Y2 VN
Z
4
0.56µF
2
– (A)
1
2
3
R2
IF V1 = A × cos (wt)
453Ω
2
C4
0.1µF
–5V
R6
2kΩ
C9
0.33µF
R5
2kΩ
RF OUT
FB
FB
+5V
–5V
C12
0.1µF
C13
0.1µF
ALL SUPPLY PINS ARE DECOUPLED AS SHOWN.
Figure 44. AGC Amplifier with 82 dB of Gain Range
−14 dB to +82 dB; the useful range is 0 dB to +82 dB if the RF
ULTRALOW NOISE AGC AMPLIFIER WITH 82 dB TO
96 dB GAIN RANGE
Figure 44 shows an implementation of an AGC amplifier with
output amplitude is controlled to 400 mV (+2 dBm). The main
82 dB of gain range using a single AD604. The signal is applied
limitation on the lower end of the signal range is the input
to connector VIN and, because the signal source is 50 Ω, a
capability of
terminating resistor (R1) of 49.9 Ω is added. The signal is then
the preamplifier. This limitation can be overcome by adding an
amplified by 14 dB (Pin FBK1 shorted to PAO1) through the
attenuator in front of the preamplifier, but that would defeat the
Channel 1 preamplifier and is further processed by the Channel 1
advantage of the ultralow noise preamplifier. It should be noted
DSX. Next, the signal is applied directly to the Channel 2 DSX. The
that the second preamplifier is not used because its ultralow
second preamplifier is powered down by connecting its COM2 pin
noise and the associated high-power consumption are overkill
after the first DSX stage. It is disabled in this application by
C1 and C2 level shift the signal from the preamplifier into the
first DSX and, at the same time, eliminate any offset contribution
of the preamplifier. C3 and C4 have the same offset cancellation
purpose for the second DSX. Each set of capacitors, combined
with the 175 Ω input resistance of the corresponding DSX,
provides a high-pass filter with a −3 dB corner frequency of
about 9.1 kHz. VOCM is decoupled to ground by a 0.1 μF
capacitor, while VREF can be externally provided; in this
application, the gain scale is set to 20 dB/V by applying 2.500 V.
Because each DSX amplifier operates from a single 5 V supply,
the output is ac-coupled via C6 and C7. The output signal can
be monitored at the connector labeled RF OUT.
connecting the COM2 pin to the positive supply. Nevertheless,
the second preamplifier can be used, if so desired, and the
useful gain range increases by 14 dB to encompass 0 dB to
96 dB of gain. For the same +2 dBm output, this allows signals
as small as −94 dBm to be measured.
To achieve the highest gains, the input signal must be band-
limited to reduce the noise; this is especially true if the second
preamplifier is used. If the maximum signal at OUT2 of the AD604
is limited to 400 mV (+2 dBm), the input signal level at the
AGC threshold is +25 μV rms (−79 dBm). The circuit as shown in
Figure 44 has about 40 MHz of noise bandwidth; the 0.8 nV/√Hz
of input referred voltage noise spectral density of the AD604
results in an rms noise of 5.05 μV in the 40 MHz bandwidth.
Rev. E | Page 19 of 32
AD604
The 50 Ω termination resistor, in parallel with the 50 Ω source
resistance of the signal generator, forms an effective resistance of
25 Ω as seen by the input of the preamplifier, creating 4.07 μV of
rms noise at a bandwidth of 40 MHz. The noise floor of this
channel is consequently 6.5 μV rms, the rms sum of these two
main noise sources. The minimum detectable signal (MDS) for
this circuit is +6.5 μV rms (−90.7 dBm). Generally, the measured
signal should be about a factor of three larger than the noise
floor, in this case 19.5 μV rms. Note that the 25 μV rms signal
that this AGC circuit can correct for is just slightly above the
MDS. Of course, the sensitivity of the input can be improved by
band-limiting the signal; if the noise bandwidth is reduced by a
factor of four to 10 MHz, the noise floor of the AGC circuit with a
50 Ω termination resistor drops to +3.25 μV rms (−96.7 dBm).
Further noise improvement can be achieved by an input matching
network or by transformer coupling of the input signal.
90
the incoming signal frequency, while passing the low frequency
AM information. The following integrator with a time constant of
2 ms set by R8 and C11 integrates the error signal presented by
the low-pass filter and changes VG until the error signal is equal
to VSET
.
For example, if the signal presented to the detector is V1 = A ×
−(V1)2/1 V. The reason for all the minus signs in the detection
circuitry is the necessity of providing negative feedback in the
control loop; actually, if VSET becomes greater than 0 V, the
control loop provides positive feedback. Squaring A × cos(ωt)
results in two terms, one at dc and one at 2ω; the following low-
pass filter passes only the −(A)2/2 dc term. This dc voltage is
now forced equal to the voltage, VSET, by the control loop. The
squarer, together with the low-pass filter, functions as a mean-
square detector. As should be evident by controlling the value of
VSET, the amplitude of the voltage V1 can be set at the input of
80
amplitude is 400 mV.
f = 1MHz
70
60
50
Figure 47 shows the control voltage, VGN, vs. the input power at
frequencies of 1 MHz (solid line) and 10 MHz (dashed line) at
an output regulated level of 2 dBm (800 mV p-p). The AGC
threshold is evident at a PIN of about −79 dBm; the highest input
power that can still be accommodated is about +3 dBm. At this
level, the output starts being distorted because of clipping in the
preamplifier.
40
30
20
10
0
–10
–20
4.5
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–30
0.1
0.5
0.9
1.3
1.7
2.1
2.5
2.9
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
VGN (V)
4
f = 1MHz
3
2
10MHz
1MHz
1
0
–1
–2
–3
–4
–80
–70
–60
–50
–40
–30
(dBm)
–20
–10
0
10
P
IN
Figure 47. Control Voltage vs. Input Power of the Circuit in Figure 44
As previously mentioned, the second preamplifier can be used
96 dB of gain and dynamic range. Because of the extremely high
gain, the bandwidth must be limited to reject some of the noise.
Furthermore, limiting the bandwidth helps suppress high-
frequency oscillations. The added components act as a low-pass
filter and dc block (C5 decouples the 2.5 V common-mode
output of the first DSX). The ferrite bead has an impedance of
about 5 Ω at 1 MHz, 30 Ω at 10 MHz, and 70 Ω at 100 MHz.
The bead, combined with R2 and C6, forms a 1 MHz low-pass
filter.
0.2
0.7
1.2
1.7
2.2
2.7
VGN (V)
The descriptions of the detector circuitry functions, comprising
a squarer, a low-pass filter, and an integrator, follow. At this
point, it is necessary to make some assumptions about the input
signal. The following explanation of the detector circuitry presumes
an amplitude modulated RF carrier where the modulating signal is
at a much lower frequency than the RF signal. The AD835
multiplier functions as the detector by squaring the output signal
presented to it by the AD604. A low-pass filter following the
squaring operation removes the RF signal component at twice
Rev. E | Page 20 of 32
AD604
At 1 MHz, the attenuation is about −0.2 dB, increasing to −6 dB
at 10 MHz and −28 dB at 100 MHz. Signals less than approximately
1 MHz are not significantly affected.
the signal amplitude compared with when they are driven
single-ended.
AD604
Figure 49 shows the control voltage vs. the input power at 1 MHz to
−95 dBm. The output signal level is set to 800 mV p-p by applying
−80 mV to the VSET connector.
1
2
3
4
5
6
7
8
9
24
23
22
–DSX1
VGN1
VREF
OUT1
VREF
C1
C2
+DSX1
PAO1
0.1µF
0.1µF
VOUT+
R1
C7
FBK1
PAI1
GND1 21
453Ω
0.1µF
VIN+
VPOS
VNEG
20
19
+5V
–5V
–5V
+5V
1
2
24
23
22
21
20
19
18
17
16
15
14
13
–DSX1
+DSX1
PAO1
FBK1
VGN1
VREF
OUT1
GND1
COM1
COM2
PAI2
VNEG 18
3
VIN–
C4
VPOS
17
C6
0.1µF
4
R2
453Ω
16
FBK2
GND2
5
PAI1 AD604 VPOS
15
VOUT–
VG
10 PAO2
OUT2
C3
0.1µF
0.1µF
6
COM1
COM2
PAI2
VNEG
VNEG
VPOS
GND2
OUT2
VOCM
VGN2
11
12
14
+DSX2
–DSX2
VOCM
7
13
VGN2
R2
499Ω
C6
560pF
C5
0.1µF
8
9
FBK2
PAO2
+DSX2
–DSX2
FB
FB
+5V
–5V
10
11
12
C3
0.1µF
C5
0.1µF
C13
0.1µF
C12
0.1µF
ALL SUPPLY PINS ARE DECOUPLED AS SHOWN.
FB
Figure 50. Ultralow Noise, Differential Input-Differential Output VGA
FAIR-RITE
Figure 51 displays the output signals VOUT+ and VOUT− after
a −20 dB attenuator formed between the 453 Ω resistors shown
plug-in. R1 and R2 are inserted to ensure a nominal load of 500 Ω
at each output. The differential gain of the circuit is set to 20 dB
by applying a control voltage, VGN, of 1 V; the gain scaling is
20 dB/V for a VREF of 2.500 V; the input frequency is 10 MHz,
and the differential input amplitude is 100 mV p-p. The resulting
differential output amplitude is 1 V p-p as can be seen on the
scope photo when reading the vertical scale as 200 mV/div.
#2643000301
Figure 48. Modifications of the AGC Amplifier to Create 96 dB of Gain Range
4.5
4.0
3.5
3.0
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2.5
2.0
1.5
1.0
0.5
0
1MHz
20mV
20ns
ACTUAL
V
OUT
+500mV
100
90
–100 –90 –80 –70 –60 –50 –40 –30 –20 –10
0
10
P
(dBm)
IN
Figure 49. Control Voltage vs. Input Power of the Circuit in Figure 48
ULTRALOW NOISE, DIFFERENTIAL INPUT-
DIFFERENTIAL OUTPUT VGA
10
–500mV
0%
Figure 50 shows how to use both preamplifiers and DSXs to
create a high impedance, differential input-differential output
VGA. This application takes advantage of the differential inputs
to the DSXs. Note that the input is not truly differential in the
sense that the common-mode voltage needs to be at ground to
achieve maximum input signal swing. This has largely to do
with the limited output swing capability of the output drivers of
the preamplifiers; they clip around 2.2 V due to having to drive
an effective load of about 30 Ω. If a different input common-mode
is recommended. The differential gain range of this circuit runs
from 6 dB to 54 dB, which is 6 dB higher than each individual
channel of the AD604 because the DSX inputs now see twice
20mV
NOTES
1. THE OUTPUT AFTER 10× ATTENUATER FORMED
BY 453Ω TOGETHER WITH 50Ω OF 7A24 PLUG-IN.
Rev. E | Page 21 of 32
AD604
The gain is controlled by means of a digital byte that is input to
The output common-mode voltage of the AD604 is set to VPOS/2
by means of an internal voltage divider. The VOCM pin is
bypassed with a 0.1 μF capacitor to ground.
MEDICAL ULTRASOUND TGC DRIVING THE
AD9050, A 10-BIT, 40 MSPS ADC
The AD604 is an ideal candidate for the time gain control (TGC)
amplifier that is required in medical ultrasound systems to limit
the dynamic range of the signal that is presented to the ADC.
in a typical medical ultrasound application.
The DSX output is optionally filtered and then buffered by
op amp output is ac-coupled into the self-biasing input of an
AD9050 ADC that is capable of outputting 10 bits at a 40 MSPS
sampling rate.
15
16
17
18
19
24
25
26
27
28
20
22
(MSB) D9
0.1µF
0.1µF
D8
D7
AD9050
1
2
24
23
22
21
20
19
18
17
16
–DSX1
+DSX1
PAO1
FBK1
PAI1
VGN1
VREF
OUT1
GND1
VPOS
VNEG
VNEG
VPOS
GND2
3
4
VREF
OUT
1kΩ
1kΩ
0.1µF
VREF
D6
3
IN
FILTER
5
D5
4
COMP
J2
A/D
OUTPUT
ANALOG
INPUT
6
D4
5
REF
BP
+5V
0.1µF
AD9631
2
3
9
AINB
D3
6
0.1µF
COM1
COM2
PAI2
–IN
+IN
6
50Ω
50Ω
10
13
14
OUT
AIN
D2
7
–5V
0.1µF
ENCODE
D1
8
OR
(LSB) D0
9
FBK2
PAO2
+DSX2
OPTIONAL
V
10
11
OUT2 15
VOCM 14
VGN2 13
0.1µF
DD
0.1µF
V
DD
12 –DSX2
0.1µF
1kΩ
AD604
CLK
100Ω
0.1µF
0.1µF
0.1µF
www.BDTIC.com/ADI
20
19
18
17
16
1
V
B
V
C
OUT
OUT
2
V
V
V
A
V
D
OUT
OUT
V
3
+15V
SSAD7226
DD
A0
A1
4
VREF
REF
5
AGND
DGND
WR 15
6
DB7
(MSB)
DB0
14
7
(LSB)
8
13
DB6
DB5
DB4
DB1
DB2 12
11
9
10
DB3
DIGITAL GAIN CONTROL
Figure 52. TGC Circuit for Medical Ultrasound Application
Rev. E | Page 22 of 32
AD604
C3
0.1µF
VG1
1
2
VGN1 24
VREF 23
OUT1 22
GND1 21
–DSX1
+DSX1
PAO1
FBK1
PAI1
VREF
OUT1
C1
C4
0.1µF
0.1µF
3
PAO1
IN1
C2
R1
NOTE 2 R2
RGN
5pF
500Ω
4
NOTE 3
C12
0.1µF
AD604
OPTIONAL
5
VPOS
VNEG
VNEG
VPOS
GND2
OUT2
VOCM
VGN2
20
19
18
17
16
15
14
13
C11
0.1µF
+5V
6
COM1
COM2
PAI2
7
–5V
C10
0.1µF
8
IN2
C9
0.1µF
NOTE 3
9
FBK2
PAO2
C8
5pF
R3
RGN
R4
500Ω
OUT2
PAO2
10
C7
0.1µF
C6
0.1µF
11 +DSX2
12 –DSX2
VOCM
VG2
0.1µF
C5
0.1µF
NOTES
1. PAO1 AND PAO2 ARE USED TO MEASURE PREAMPS.
2. RGN = 0 NOMINALLY; PREAMP GAIN = 5, RGN = OPEN; PREAMP GAIN = 10.
3. WHEN MEASURING BW WITH 50Ω SPECTRUM ANALYZER, USE 450Ω IN SERIES.
Figure 53. Basic Test Board
HP3577B
OUT
R
A
HP11636B
POWER
SPLITTER
www.BDTIC.com/ADI
50Ω
0.1µF
450Ω
PAI
AD604
DUT
49.9Ω
Figure 54. Setup for Gain Measurements
Rev. E | Page 23 of 32
AD604
EVALUATION BOARD
Figure 55 is a photograph of the AD604 evaluation board assembly.
Multiple input connections, test points, jumper selectable options,
and on-board trims offer convenience when configuring the
AD604 in various operating modes.
The evaluation board requires only a dual 5 V supply capable of
200 mA or higher to operate both channels. Prior to shipment,
the evaluation board is fully tested. Users need only attach
power supply leads and the appropriate test equipment to
the board.
Because of this flexibility, not all component positions on the
board are populated when the board is shipped. Installing or
changing additional parts is optional.
The AD604-EVALZ is fabricated on a 4-layer board with inner
power and ground layers. The AD604 is a stable, trouble-free
device; however, as with all high-frequency integrated circuits,
power and ground planes help to ensure consistency in
performance.
Figure 56. AD604 Evaluation Board—Component Side Silk Screen
DSX INPUT CONNECTIONS
The DSX inputs can be connected in single-ended or differential
www.BDTIC.com/ADI
configurations. SMA connectors are provided for each of the
inputs and are labeled CHx VGA IN (+) and CHx VGA IN (−).
JP6 and JP15 select between the preamplifier outputs and the
DSX inputs.
For direct drive of the Channel 1 VGA, insert a jumper in the
top position of JP6. For direct drive of the Channel 2 VGA,
insert a jumper in JP14 and verify that there are no jumpers in
circuit details.
Differential DSX Inputs
Differential inputs are possible using both polarities of the
VGA SMA connectors and appropriate jumpers. Inserting a
jumper in the lower position of JP5 selects the negative input
Figure 55. AD604 Evaluation Board Assembly
of Channel 1. A jumper in the top position of JP6 selects the
USING THE PREAMPLIFIER
positive input of Channel 1. A jumper in the JP16 rightmost
position selects the negative input of Channel 2, and a jumper
in JP14 selects the positive input. Verify that there are no jumpers
in JP15 or JP13.
To use the preamplifiers, simply connect a signal source to CH1
PREAMP IN and/or CH2 PREAMP IN via the SMA connectors.
terminated with 50 Ω resistors at locations R7 and R8.
Because the VGA section of the AD604 uses a single 5 V supply,
the DSX inputs are ac-coupled. Decoupling capacitors are provided
on the evaluation board.
To enable the preamplifiers, insert jumpers in the JP8 and JP9
rightmost positions; this connects COM1 and COM2 to ground.
Power down the preamplifiers by inserting jumpers in the JP8
and JP9 leftmost positions.
The DSX input impedance is approximately 200 Ω. Optional
66.5 Ω resistors can be installed across the inputs at positions
R5, R6, R9, and R10 to establish a 50 Ω terminating load.
Rev. E | Page 24 of 32
AD604
Connecting the DSX Inputs to the Preamplifiers
OUTPUTS
To connect the DSX inputs to the preamplifiers, install jumpers
in the JP6 lower position and in JP15. Verify that the jumpers in
JP13 and JP14 are removed.
The DSX outputs are available on OUT1 and OUT2 SMA
connectors and are series terminated with decoupling capacitors
and 49.9 Ω series resistors. These components can be replaced
to accommodate other output impedances.
Cascaded DSX
DC OPERATING CONDITIONS
To channel-cascade the two channels, insert a jumper in JP13.
The resulting single-channel gain range is 96 dB. Verify that
JP14 and JP15 are removed.
convenient dc level adjustments of gain, reference voltage,
and their functions.
The gains of cascaded VGAs can be controlled independently
or in common. For common control, insert a jumper in the top
position of JP4. To use the trimmer as a gain control, insert a
jumper in JP1. For external control, remove JP1 and connect a
signal source at VGN1 or VGN2 test loop.
Table 4. Trimmer Functions
Trimmer
Function
R1
R2
R3
R4
Gain of Channel 1
PREAMPLIFIER GAIN
Reference voltage adjustment
Output common-mode voltage adjustment
Channel 2 gain adjustment
Jumpers in JP7 and JP12 select between two preamplifier gains:
14 dB and 20 dB. Intermediate gains are derived by installing
resistors in the R11 and R12 positions. The 14 dB and 20 dB
preset gains are accurate due to close matching of thin film
resistors. The gain accuracy after installing external resistors is
subject to inherent tolerance of absolute accuracy.
Table 5. Jumpers
Jumper No.
Function
1
2
3
4
5
6
7
8
9
12
13
14
15
16
Connects R1 gain adjust wiper to VGN1.
Connects R2 reference voltage trimmer to VREF input.
www.BDTIC.com/ADI
Connects common-mode voltage trimmer to VOCM.
Connects VGN2 to R4 Channel 2 gain trimmer or to VGN1 or common gain adjustment.
Connects –DSX1 to CH1 VGA IN (−) or to ground.
Connects +DSX1 (ac-coupled) to preamplifier output of Channel 1 or to the CH 1 VGA IN (+) SMA connector.
When open, the Preamp 1 gain is 20 dB; Preamp 1 gain is 14 dB when a shunt is installed.
Shunt in left position disables Preamp 1; shunt in rightmost position enables Preamp 1.
Shunt in left position disables Preamp 2; shunt in rightmost position enables Preamp 2.
When open, the Preamp 2 gain is 20 dB; Preamp 2 gain is 14 dB when a shunt is installed.
Cascades DSX2 with DSX 1 when a jumper is inserted.
Connects +DSX2 (ac-coupled) to preamplifier output of Channel 2 or to the CH 2 VGA IN (+) SMA connector.
Connects +DSX2 (ac-coupled) to preamplifier output of Channel 2.
Connects –DSX2 to CH2 VGA IN (−) or to ground.
Rev. E | Page 25 of 32
AD604
+5V
J1
GND1 GND2 GND3 GND4
CH1 VGA
IN (–)
–DSX1
GN1
ADJ
R1
10kΩ
C8
JP5
0.1µF
A
VREF
JP1
R5
+5V
B
AD604
VGN1
JP2
J2
U1
VREF
ADJ
R2
10kΩ
CH1 VGA
IN (+)
+DSX1
1
2
3
4
5
6
24
23
22
21
20
19
18
17
16
15
14
13
–DSX1
+DSX1
PAO1
FBK1
PAI1
VGN1
VREF
OUT1
C9
JP6
C5
0.1µF
1nF
A
C13
0.1µF
R6
C6
0.1µF
B
PAO1
OUT1
+5V
J3
OUT1
R13
49.9Ω
R11
JP8
JP7
J5
CH 1 PREAMP
IN
GND1
VPOS
PAI1
+5V
+5V
GND
–5V
–5V
R7
49.9Ω
COM1
COM2
PAI2
VNEG
VNEG
VPOS
GND2
OUT2
VOCM
VGN2
C4
0.1µF
C3
10µF
10V
JP9
+
J6
CH 2 PREAMP
IN
7
8
PAI2
C1
10µF
10V
+
C2
0.1µF
R8
49.9Ω
JP12
PAO2
9
FBK2
PAO2
+DSX2
–DSX2
C14
0.1µF
R14
49.9Ω
R12
OUT2
JP15
J7
10
J4
OUT2
CH2 VGA
IN (+)
+DSX2
VOCM
JP14
11
12
+5V
C7
0.1µF
JP13
C10
0.1µF
R9
JP3
R3 VOCM
1kΩ ADJ
J8
CH1 VGA
IN (–)
–DSX2
JP16
C11
0.1µF
VGN2
JP4
+5V
A
B
A
B
GN2
ADJ
R10
R4
C12
1nF
1kΩ
NOTES
1. PARTS IN GRAY ARE NOT INSTALLED.
Figure 61. Evaluation Board Schematic
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Table 6. Bill of Materials
Qty. Type
Description
Reference Designator
Manufacturer
Part Number
TP-104-01-02
TP-104-01-06
TP-104-01-00
TP-104-01-07
1
1
5
14
Test loop
Red
Blue
Black
Purple
+5 V
−5 V
Components Corp.
Components Corp.
Components Corp.
Test loop
Test loop
Test loop
GND, GND1, GND2, GND3, GND4
+DSX1, +DSX2, −DSX1, −DSX2, OUT1, OUT2, PAI1, Components Corp.
PAI2, PAO1, PAO2, VGN1, VGN2, VOCM, VREF
2
10
2
8
8
6
4
4
Capacitor
Capacitor
Capacitor
Tantalum 10 μF, 10 V, A size C1, C3
Nichicon
Panasonic
Panasonic
Amphenol
Berg
Molex
Bourns
Panasonic
F931A106MAA
PCC1840CT-ND
ECU-V1H102KBN
901-143-6RFX
69157-2
22-11-2032
3361P-1-103G
ERJ-6ENF49R9
0.1 μF, 50 V, 20%, 0805
C2, C4, C6, C7, C8, C9, C10, C11, C13, C14
SM, 1000 pF, 50 V, 0805
C5, C12
Connector SMA FEM PC Mount, RA
J1, J2, J3, J4, J5, J6, J7, J8
JP1, JP2, JP3, JP7, JP12, JP13, JP14, JP15
JP4, JP5, JP6, JP8, JP9, JP16
R1, R2, R3, R4
Header
Header
Trimmer
Resistor
0.1”center 2-pin
0.1”center 3-pin
10 kΩ, 1/4" SM
49.9 Ω, 1%, 1/10 W, 0805
R7, R8, R13, R14
1
Integrated 40 MHz dual low
U1
Analog Devices, Inc. AD604AR
circuit
noise VGA
10
Jumper
Mini jumper; install in
headers at JP1, JP2, JP3,
JP4 lower, JP5 lower, JP6
lower, JP8 right, JP9 right,
JP15, JP16 left
Rev. E | Page 27 of 32
AD604
OUTLINE DIMENSIONS
1.280 (32.51)
1.250 (31.75)
1.230 (31.24)
24
1
13
12
0.280 (7.11)
0.250 (6.35)
0.240 (6.10)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.100 (2.54)
BSC
0.060 (1.52)
MAX
0.195 (4.95)
0.130 (3.30)
0.115 (2.92)
0.210 (5.33)
MAX
0.015
(0.38)
MIN
0.150 (3.81)
0.130 (3.30)
0.115 (2.92)
0.015 (0.38)
GAUGE
0.014 (0.36)
0.010 (0.25)
0.008 (0.20)
PLANE
SEATING
PLANE
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
0.430 (10.92)
MAX
0.005 (0.13)
MIN
0.070 (1.78)
0.060 (1.52)
0.045 (1.14)
COMPLIANT TO JEDEC STANDARDS MS-001
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
Figure 62. 24-Lead Plastic Dual In-Line Package [PDIP]
Narrow Body
(N-24-1)
Dimensions shown in inches and (millimeters)
www.BDTIC.com/ADI
15.60 (0.6142)
15.20 (0.5984)
24
13
7.60 (0.2992)
7.40 (0.2913)
1
10.65 (0.4193)
10.00 (0.3937)
12
0.75 (0.0295)
0.25 (0.0098)
45°
2.65 (0.1043)
2.35 (0.0925)
0.30 (0.0118)
0.10 (0.0039)
8°
0°
COPLANARITY
0.10
SEATING
PLANE
0.51 (0.0201)
0.31 (0.0122)
1.27 (0.0500)
BSC
1.27 (0.0500)
0.40 (0.0157)
0.33 (0.0130)
0.20 (0.0079)
COMPLIANT TO JEDEC STANDARDS MS-013-AD
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 63. 24-Lead Standard Small Outline Package [SOIC_W]
Wide Body
(RW-24)
Dimensions shown in millimeters and (inches)
Rev. E | Page 28 of 32
AD604
8.50
8.20
7.90
13
12
24
5.60
5.30
5.00
8.20
7.80
7.40
1
0.25
0.09
1.85
1.75
1.65
2.00 MAX
8°
4°
0°
0.95
0.75
0.55
0.38
0.22
0.05 MIN
SEATING
PLANE
COPLANARITY
0.10
0.65 BSC
COMPLIANT TO JEDEC STANDARDS MO-150-AG
Figure 64. 24-Lead Shrink Small Outline Package [SSOP]
(RS-24)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD604AN
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
Package Option
24-Lead Plastic Dual In-Line Package [PDIP]
24-Lead Plastic Dual In-Line Package [PDIP]
24-Lead Standard Small Outline Package [SOIC_W]
24-Lead Standard Small Outline Package [SOIC_W]
24-Lead Standard Small Outline Package [SOIC_W]
24-Lead Standard Small Outline Package [SOIC_W]
24-Lead Shrink Small Outline Package [SSOP]
24-Lead Shrink Small Outline Package [SSOP]
24-Lead Shrink Small Outline Package [SSOP]
24-Lead Shrink Small Outline Package [SSOP]
24-Lead Shrink Small Outline Package [SSOP]
24-Lead Shrink Small Outline Package [SSOP]
Evaluation Board
N-24-1
N-24-1
RW-24
RW-24
RW-24
RW-24
RS-24
RS-24
RS-24
RS-24
RS-24
RS-24
AD604AR
AD604AR-REEL
AD604ARS
AD604ARS-REEL
AD604ARS-REEL7
AD604ARSZ-RL1
1 Z = RoHS Compliant Part.
Rev. E | Page 29 of 32
AD604
NOTES
www.BDTIC.com/ADI
Rev. E | Page 30 of 32
AD604
NOTES
www.BDTIC.com/ADI
Rev. E | Page 31 of 32
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